Analog Devices AD602 manual U1 AD600

AD600/AD602

INPUT	0dB
	CAL
1V RMS

MAX

(SINE WAVE)	R1
	115Ω
	R2 200Ω
	R3
	133kΩ
	U3A
	1/2
	AD712
	VG
	15.625mV/dB

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

R4

3.01kΩ

				VRMS
				AF/RF
				OUTPUT
C1						C4
0.1∝F						4.7∝F					+6V
										+6V DEC
		C1HI									FB
1	16				1	VINP	VPOS	14			0.1∝F
2	15	A1CM		NC	2	U2		13	NC	+6V DEC
						AD636
A1		A1OP
	14				3			12
3				–6V DEC		VNEG			NC	–6V DEC
4	13	VPOS	+6V DEC		4	CAVG		11	NC		0.1∝F
										R7	FB
REF		VNEG	C2
	12								R6	56.2kΩ
5			–6V DEC 2∝F	NC	5	VLOG	COMM	10
		A2OP							3.16kΩ		–6V
6	11			NC	6	BFOP	LDLO	9
										POWER SUPPLY
A2		A2CM
	10									DECOUPLING NETWORK
7					7	BFIN	VRMS	8
8	9	C2HI								+316.2mV
U1 AD600								1/2	U3B	C3
								AD712
R5										1∝F

16.2kΩ

VOUT

+100mV/dB

0V = 0dB (AT 10mV RMS)

NC = NO CONNECT

Figure 19. The Output of This Three-IC Circuit Is Proportional to the Decibel Value of the RMS Input

Note that the peak “log output” of ± 4 V requires the use of

±6 V supplies for the dual op amp U3 (AD712) although lower supplies would suffice for the AD600 and AD636. If only ± 5 V supplies are available, it will be either necessary to use a reduced

value for VSCALE (say 1 V, in which case the peak output would be only ± 2 V) or restrict the dynamic range of the signal to

about 60 dB.

As in the previous case, the two amplifiers of the AD600 are used in cascade. However, the 6 dB attenuator and low-pass fil- ter found in Figure 1 are replaced by a unity gain buffer ampli- fier U3A, whose 4 MHz bandwidth eliminates the risk of instability at the highest gains. The buffer also allows the use of a high impedance coupling network (C1/R3) which introduces a high-pass corner at about 12 Hz. An input attenuator of 10 dB (X0.316) is now provided by R1 + R2 operating in conjunction with the AD600’s input resistance of 100 Ω. The adjustment provides exact calibration of the logarithmic intercept VREF in critical applications, but R1 and R2 may be replaced by a fixed resistor of 215 Ω if very close calibration is not needed, since the input resistance of the AD600 (and all other key parameters of it and the AD636) are already laser trimmed for accurate opera- tion. This attenuator allows inputs as large as ± 4 V to be ac- cepted, that is, signals with an rms value of 1 V combined with a crest factor of up to 4.

The output of A2 is ac coupled via another 12 Hz high-pass fil- ter formed by C2 and the 6.7 kΩ input resistance of the AD636. The averaging time constant for the rms-dc converter is deter- mined by C4. The unbuffered output of the AD636 (at Pin 8) is compared with a fixed voltage of +316 mV set by the positive

supply voltage of +6 V and resistors R6 and R7. (VREF is pro- portional to this voltage, and systems requiring greater calibra- tion accuracy should replace the supply dependent reference with a more stable source.)

Any difference in these voltages is integrated by the op amp U3B, with a time constant of 3 ms formed by the parallel sum of R6/R7 and C3. Now, if the output of the AD600 is too high, V rms will be greater than the “setpoint” of 316 mV, causing the output of U3B—that is, VOUT—to ramp up (note that the inte- grator is noninverting). A fraction of VOUT is connected to the inverting gain-control inputs of the AD600, so causing the gain to be reduced, as required, until V rms is exactly equal to

316 mV, at which time the ac voltage at the output of A2 is forced to be exactly 316 mV rms. This fraction is set by R4 and R5 such that a 15.625 mV change in the control voltages of A1 and A2—which would change the gain of the cascaded amplifi- ers by 1 dB—requires a change of 100 mV at VOUT. Notice here that since A2 is forced to operate at an output level well below its capacity, waveforms of high crest factor can be tolerated throughout the amplifier.

To check the operation, assume an input of 10 mV rms is ap- plied to the input, which results in a voltage of 3.16 mV rms at the input to A1, due to the 10 dB loss in the attenuator. If the system operates as claimed, VOUT (and hence VG) should be zero. This being the case, the gain of both A1 and A2 will be

20 dB and the output of the AD600 will therefore be 100 times (40 dB) greater than its input, which evaluates to 316 mV rms, the input required at the AD636 to balance the loop. Finally, note that unlike most AGC circuits, needing strong temperature compensation for the internal “kT/q” scaling, these voltages, and thus the output of this measurement system, are tempera- ture stable, arising directly from the fundamental and exact exponential attenuation of the ladder networks in the AD600.

Typical results are presented for a sine wave input at 100 kHz. Figure 20 shows that the output is held very close to the setpoint of 316 mV rms over an input range in excess of 80 dB.

–12– REV. A