SK201B is a 13VDC signal trigger output which is active whenever the amplifier is powered up. R218 and DZ207 / C223 provide a reference voltage which is buffered by TR200. TR201 and R217 act as a current limit and prevent damage due to a short circuit on the output of SK201B. The maximum current is approximately 65mA.
TR203 and TR202 are a complementary Darlington pair which turn on mains relay RLY200 when activated by a signal from the microprocessor.
TR204 and its associated components are to detect whenever AC mains is present at the IEC socket. This is to notify the microprocessor if the user has unplugged the mains cord, so that it can take the necessary action (muting all the outputs and switching off the mains relay). The reservoir capacitors should last at least 4 mains cycles which gives the microprocessor plenty of time for a controlled shutdown.
TR204 forms a monostable circuit. Each cycle of AC turns on TR204 via R211. TR204 then ‘shunts’ C229 ensuring that it is kept at a low potential. If more than one mains cycle is missing, then R219 charges up C229 sufficiently to trigger Schmitt inverter IC202E thus passing on a logic signal to the microprocessor. The use of a Schmitt inverter for IC202 is to ensure that the micro receives ‘clean’ logic levels - the hysteresis voltage (about 0.5V) is sufficient to prevent circuit noise from producing a string of ‘ghost’ signals when analogue levels are near the threshold point.
TH200 is a positive tempco thermistor placed adjacent to the heatsink on which the output transistors are mounted. When the temperature of the thermistor exceeds 90 degrees Celsius the thermistor goes to a high impedance and so the input to IC202F goes low. This triggers a HIGH output to the micro indicating thermal overload.
The VI protection signals from the left and right channels pass into IC202A and IC202B respectively, to be ‘cleaned up’ via the Schmitt trigger. They are then NOR’d using TR205 which sends a HIGH signal to the micro in the event of either channel suffering a short circuit or current overload. Exactly the same approach is used for the DC fault lines using IC202C and IC202D.
L882 Circuit Sheet 3
This is the main audio power amplifier circuit. The amplifier is a class B design, which uses SAP ‘audio’ transistors in a symmetrical current feedback configuration. Input and feedback paths are DC coupled and there is an active integrating servo to remove DC offsets from the output.
The basic principle of operation is follows:
The input signal is amplified by a factor of 2 in IC300A. This drives a 44 impedance to ground causing the supply pin currents to change with the signal level. These changing supply pin currents are then ‘reflected’ by a pair of complementary Wilson mirrors and passed on to a series of buffer transistors before being connected to the load. The ‘feedback current’ flows back from the output terminal via R331 and R332 and attempts to provide the current necessary to allow IC300A to swing its output without drawing excessive current from its supply pins, thus making the change in supply current very small indeed. This is why the term ‘current feedback’ is used - it is the current flowing in the feedback resistors that sets the overall gain of the amplifier.
IC300B acts as an inverting integrator and its purpose is to remove DC from the loudspeaker output. Any positive DC offset will cause the output of IC300B to go negative, thus increasing the current in its negative supply pin and pulling the output voltage back towards zero. R330 and C317 set the time constant of this integrator (0.47 seconds) so that audio frequency components are ignored and only DC and subsonic frequencies are removed.
The input to the amplifier is limited to ±5.4V via back-to-back zener diodes DZ302 and DZ303. This is to prevent the user from grossly overdriving the input to the amplifier and possibly causing damage. The diodes appear before series resistor R324 so that their variable capacitance does not introduce high frequency harmonic distortion.
R324, R327 and C316 act as an input filter - this is a first order low pass filter with a corner frequency of around 340kHz to prevent RF signals from being injected into the front end of the amplifier. The corner frequency was chosen such that the phase shift introduced is less than 5 at 20kHz (considered by the AES to be the minimum perceptible relative amount by the human ear). The input impedance of the amplifier is 23kW at DC, falling to around 14kW at 20kHz.
Operational amplifier IC300A is acting as a non-inverting gain of 2, driving the input signal into a 44W impedance to ground via R322 and R337. Its output voltage will be an accurate amplification of its input voltage (i.e. the signal on pin 1 should look identical to that on pin 3 but at twice the amplitude). The op- amp is used in a slightly unusual configuration here, in that its power supply pins are used as a (current) output, and its output pin is used as a (current) feedback.
Transistors TR311 and TR303 supply the ±15V rails to the op- amp, and act as cascades to pass its supply pin currents through to the current mirrors, which sit at a potential too high for the op-amp to be connected directly.
TR300, TR301 and TR321 form a PNP Wilson current mirror, which reflects the current sunk by the positive supply pin of IC300. Likewise TR314, TR315 and TR320 form an NPN Wilson current mirror, which reflects the current sourced by the negative supply pin of IC300.
R315 thru R318 provide emitter degeneration of approximately 300mV for the current mirrors (as they pass about 3mA DC in quiescent conditions), to ensure accurate operation independent of the small variations between the transistors in the current mirrors. They also ensure that the current passing down the next stage is reasonably constant as the internal temperature of the amplifier changes, swamping out small thermal variations in the VBE of the mirror transistors.
R319 and R320 slightly decouple the rails to the current mirrors from the main power rails of the amplifier, to allow the bootstrap circuit to operate. The bootstrap consists of C302 and C306 with metal film power resistors R352 and R353. The bootstrap is provided to allow the power supply rails of the current mirrors to go up and down slightly with the output signal into the loudspeaker. This enables the driver stage to fully saturate the output transistors and thus give the greatest power output and best thermal efficiency for any given power rail voltage. The voltage on the ‘inside’ end of R319 and R320 will vary by about 12 volts peak to peak at full output power, rising above the main power rails during signal peaks.
C307 and C308 with R333 and R335 provide the compensation necessary to ensure stability when the loop is closed. They are Miller capacitors which dramatically reduce the transimpedance (i.e. current to voltage gain) of the current mirrors at high frequencies. The present value of 47pF provides for a unity gain open loop bandwidth of around 75MHz, whilst ensuring a closed loop gain margin of around 6dB (note that gain margin in a current feedback design is not dependent on system bandwidth to a first order approximation). R333 and R335 provide a ‘zero’ in the open loop frequency response which is tailored to give the best time domain performance (i.e. to make high frequency square waves look square with minimal ringing or overshoot).
DZ304 and C311 provide a fixed 4.7V bias voltage to allow the following stages to operate correctly. C311 is there to ensure that
